1. Field of the Invention
This invention relates to a new class of phase lock loop (PLL) frequency synthesizers utilizing very high reference clock frequency (in the order of 100 MHz, but not limited to) and very high comparison frequencies (of the same order as the reference clock frequency), but providing fine frequency resolution, where frequency steps increments can be orders of magnitude smaller (in the order of 10 kHz, but not limited to) than the comparison frequency, and especially to their use as single loop PLL frequency synthesizers having very low phase noise, thus offering significant performance improvements for advantageous applications such as cable television (CATV), high speed digital communications (digital TV and high speed cable data modems for internet applications), wireless communications and other consumer and commercial electronics devices in high frequency (HF), very high frequency (VHF) and ultra high frequency (UHF) bands and beyond, but which are simple and inexpensive to manufacture, and suitable for integration into modern integrated circuits, as a single-chip solution. In addition, this invention relates to the field of precision phase control of carriers in coherent multi-carrier transmission systems.
2. Background of the Related Art
Modern communication and other electronics systems often require generation of internal signals necessary for various processing functions, ranging from local oscillator sources for signal frequency up or down conversion or modulation/demodulation purposes, to various clock signals used by processors and controllers. Although the requirements regarding the quality and speed at which these signals are generated highly depend on the specific purity, and in the case of commonly used frequency agile systems in communications, the tuning speed at which the channel frequencies can be dynamically change may be important as well. As is well known in the art, it is the goal of frequency synthesizers to accomplish the above tasks. It is therefore a general requirement for synthesizers to be able, by using reference signal or signals, to generate spectrally clean signals (or stabilize other signals) at required frequencies, having spectral purity, phase noise and tuning speed characteristics that are consistent with the system and application requirements. Furthermore, it is the synthesizers that are often called upon to reject (i.e. attenuate) undesired components that may accompany a signal that is being synthesized, stabilized or synchronized. An example of such a case is when a synthesizer is used to reduce the relatively high close-in phase noise inherently present in voltage controlled oscillators (VCO). In addition, it is often desired, if not required that the synthesizer can perform (or assist in) phase or frequency modulation or demodulation. In specific applications utilizing large number of coherent carriers carried over various mediums, such as in fiber optic links and coaxial cable distribution plant used in cable television, precision phase control of the carriers may be desired for the purpose of improving distortion performance of the transmission system, and in such cases it would be advantageous if frequency synthesizer could accomplish the carrier phase control function as well.
In modern communication systems deploying phase or frequency modulation, such as quadrature amplitude modulation (QAM) used in digital cable TV and high speed data modems, quadrature phase modulation (QPSK) used in digital cellular telephony, FM modulation used in analog cellular telephony, and many other communication systems employing similar modulation formats, it is the phase noise performance of the synthesized signals that is becoming increasingly important with a specific reason to preserve system performance and prevent degradation. As is well known in the art, excessive phase noise of the sources used to process modulated signals that contain information in it's phase or frequency can cause degradation in signal to noise ratio (S/N) or bit error rate (BER) of the desired signal, resulting in reduced sensitivity and unsatisfactory system performance. This is particularly true in the systems utilizing narrow band transmission, such as lower deviation FM or lower symbol rate QPSK systems, where phase noise of local oscillators (LO) used to down convert RF signals to IF frequencies, or vice cersa, is effectively added to the demodulated signal carrying information, thus directly degrading the S/N or BER. In the systems with closely spaced channels, it is also possible that the LO phase noise can cause degradation in sensitivity due to adjacent channel interference, whereby the LO phase noise side bands can in the receiver down-convert the adjacent channel and place it directly on the desired signal and so cause interference, or in the case of a transmitter LO, the transmitted LO phase noise side band occurring at the adjacent channel frequency can cause direct interference in the receivers tuned to that adjacent channel frequency.
When discussing the phase noise, it is important to note that phase noise is random in nature, having continuous spectral distribution around the synthesized signal. It should be distinguished from discrete undesired sidebands that may also exist as various spurious components in the vicinity of the signal. As is well known in the art, the discrete spurious sidebands of the synthesized signal are none less important than the random phase noise sidebands, since they can cause similar, and in many cases even worse undesired effects. The burden is usually on the synthesizer to ensure that all the above requirements are met, emphasizing that the spectral purity and phase noise requirements are most often the governing factor in the system design and implementation choices. Finally, it is always desirable that the above functions are realized in an inexpensive, repeatable and easily manufacturable way.
There are two well known general techniques for implementing frequency synthesizers—direct synthesis and indirect synthesis methods. Various forms of the combination of these two techniques are also well known in the art. The direct methods include direct analog synthesis (DAS) and direct digital synthesis (DDS). The indirect methods primarily rely on utilization of phase lock loops (PLL).
The direct analog synthesis method generates the desired output frequency or frequencies by the combination of such means as analog up/down conversion, multiplication, filtering and attenuation/amplification of one or more reference signals. Frequency division may also be utilized, whereby digital dividers are primarily used.
The direct digital synthesis utilizes a sampled process, where digital words representing desired waveforms are computed and converted with the aid of a sampling clock to a desired analog signal in digital to analog converter followed by antialiasing filter.
The indirect synthesis technique utilizes a method where one oscillator (in single loop PLLs) or more oscillators (in multiple loop PLI,s) are forced in a controlled way to be phase (and frequency) locked to one or more reference signals. While the phase locked loops are most commonly used to accomplish this task, other means, such as injection-locked oscillators (also known as synchronous oscillators) may also be used for this function.
Each of the above techniques has its advantages and disadvantages regarding frequency coverage, frequency agility and switching speed, phase noise and spurious sideband performance, power consumption, size, cost, manufacturability, repeatability etc. Many of these factors are contradictory in nature to each other, and it is the matter of intended application that will govern the considerations and choices as to which method or methods is the most suitable to use. Although the characteristics of each of the techniques regarding these factors are well known in the art, for the purpose of completeness of the discussion, the key characteristics for each of the methods is summarized below:
DAS—possibly the best spectral purity (phase noise, close-in spurs) and fastest switching time attainable, wide frequency coverage but limited frequency agility, larger size, higher complexity and cost, may require adjustments in manufacturing—in general not suitable for large volume, low cost applications.
DDS—very good noise and close-in spectral purity performance, fast switching with continuous phase capability, excellent agility but limited frequency coverage (currently at about 100 MHz practical implementation limit), medium to high cost (increases with frequency), could be used in limited number of high volume applications.
PLL based frequency synthesizers—moderate to good noise and close—in spur performance, slow to medium switching speed, wide frequency coverage and agility (usually limited by VCO), lowest to medium cost, moderate to good repeatability, low power.
The single loop PLL based frequency synthesizers are by far the most widely utilized type in large volume communications systems applications, since in most cases they offer the best compromise of all of the above competing factors (i.e. low cost, simple construction with low parts count and low power consumption, acceptable performance levels). We will hereinafter in the background discussion focus on the single loop PLL type of synthesizers, with the emphasis on the shortcomings in the prior art of this type, as some of these shortcomings are the subjects being addressed by the present invention.
PLL synthesizers, as mentioned earlier, operate on the principle of phase locking an oscillator (or multiplicity of) to a reference signal (or signals). To simplify the following description, we will narrow the discussion (but not limit to) a case involving only one oscillator and one reference signal source.
In general, every PLL represents a closed loop negative feedback control system, where the controlling variable is the phase of the controlled element, and a controlled element is an oscillator with variable frequency capability, most often of the VCO type. Such a feedback system requires a mechanism to generate an error signal, upon comparing the controlling variable (oscillator's phase) with the equivalent parameter of the reference (reference's phase). This error signal is by the means of a loop filter converted into a control signal, which is in turn used to steer the oscillator in the direction that reduces the error signal towards zero or some finite value.
In a PLL, the error signal is generated by the phase comparator (phase detector). The phase detector compares the phase of the oscillator with that of the reference signal, and provides an error signal proportional to the phase difference. The error signal is further conditioned by the loop filter. The loop filter usually consists of either a passive or active single or multiple pole filter, which may be followed with an amplifier, to produce a control signal. Applying this control signal from the output of the loop filter to the tuning line of the oscillator, closes the feedback loop, and completes the PLL circuit. However, before a phase lock can occur, a frequency lock must occur. During acquisition of phase lock, depending upon the amount of frequency difference between the reference and oscillator signals, the phase detector alone may not provide an adequate steering signal that will guide the frequency of the oscillator in the right direction, and for the purpose of assisting, or enabling acquisition, the so called phase-frequency detectors (PFD), or other means such as frequency pre-tuning or frequency sweeping are utilized. In the acquisition stage, the PFD (or above mentioned other means) provides a DC steering signal of the right polarity, consistent with the sense of the frequency difference, which guides the oscillator in the correct direction towards a lock.
The ability of the loop to correct any fluctuations of the phase that the oscillator may have depends upon a loop bandwidth (LBW) of the loop. Within the LBW, the loop will substantially eliminate (i.e. reject) any phase fluctuations (phase noise) that the oscillator itself may have, while the oscillator will “copy”, i.e. track the phase of the reference. It will also track any noise that falls within the LBW generated elsewhere in the loop (e.g. the noise of the phase detector). This noise is added to the phase noise of the reference signal. The loop will multiply this combined noise by a factor which equals to Ntot (total division ratio in the loop), and then phase modulate the VCO with this multiplied noise. As will be shown later, this noise multiplication is often the key factor causing degradation of phase noise performance in synthesizers.
Typically, the desired output frequency of the oscillator differs from that of the reference frequency. Furthermore, in agile systems the oscillator needs to be tunable to different frequencies at different times. For this purpose, some sort of programmable frequency scaling is required. This function is accomplished by the means of so called prescalers (either of a fixed or of a variable division ratio, or modulus) and programmable counters. Since programmable counters have limitations in terms of the upper frequency limit where they can operate, often prescalers are used to scale the oscillator frequency down and bring it within the operational frequency range of the programmable counters.
Often it is necessary to scale down the reference frequency as well, so that the reference frequency is divided down to a lower frequency, referred to as a comparison frequency, which is the frequency at which the comparison of the phase (and frequency) of the scaled versions of both the oscillator and reference signals actually occurs. In the lock conditions, the scaled frequencies of the oscillator and reference signals are made equal by the PLL action, and the frequency relationships are expressed with the following equations:                     fc        =                  fr          R                                    (        2        )            where fosc is the oscillator output frequency and Ntot is the total division ratio of the oscillator signal, ft is the reference frequency, R is the division ratio of the reference frequency, and fc is the comparison frequency.
As is well documented in the literature related to the art, the primary cause of phase noise at the output of the synthesized oscillators s the combination of the residual VCO phase noise (the portion of the oscillator noise that is not rejected by the loop), and the noise power of the loop components (such as dividers and phase detectors). This loop noise is multiplied by the loop by a factor of Ntot2, or 20·log(Ntot). Any random noise and discrete spurs occurring at the output of the phase comparator within the LBW will be multiplied by this factor and then applied to the oscillator where the phase modulation of the oscillator by these signals will occur. Outside of the LBW, the loop will attenuate such signals according to the transfer function of the loop. In the case of relatively low comparison frequency fc and relatively high oscillator frequency fosc, the 20·log(Ntot) figure can amount to a very large quantity, exceeding the residual VCO phase noise and thus becoming a dominant and limiting factor in the ultimate noise performance achievable in the system.
One example of adverse effects of noise multiplication (or noise gain) in the loop is in the cable television systems, where channel frequency spacing is 6 MHz, but the channel carrier frequencies must be integer multiples of 12.5 kHz (in a standard frequency plan, most channel frequencies are multiples of 0.25 MHz, but due to regulatory requirements by Federal Communication Commission—FCC, a few channels must be offset from this frequency grid by 25 kHz, and some other channels by 12.5 kHz; furthermore, some cable TV systems use a frequency plan where all channels are offset by 12.5 kHz from a standard plan—the so called “incrementally related carriers”—IRC plan). To accommodate these channel frequency requirements, a comparison frequency of 12.5 kHz is required. For a local oscillator operating around 970 MHz in a cable head-end modulator, or in a set-top converter, the noise multiplication figure will amount to 98 dB. Another example of noise multiplication is in an oscillator, synthesized at about 900 MHz, with step size of 30 kHz, common in cellular telephony. This step size will demand a comparison frequency of 30 kHz, and the noise multiplication figure in this case will be close to 90 dB. The noise gain of this magnitude can place severe limitations as to the achievable phase noise performance in the system, such as in the above two examples.
By examination of equation (1) and (2), it becomes evident that the frequency step resolution, i.e. the minimum achievable frequency increment of the oscillator frequency is the function of both the comparison frequency fc and the minimum available increment of the division ratio Ntot, as expressed in the following equation:Δfosc=fstep=ΔNtot·fc  (3)where Δfosc=fstep is the minimum achievable frequency increment of the oscillator frequency and ΔNtot is the minimum available increment of the total divider count Ntot. By examining equation (3), it becomes clear that if Ntot is an integer (and therefore having a minimum increment of ΔNtot=1) the smallest achievable step size Δfosc=fstep is equal to fc. Furthermore, from equation (1) it follows that Ntot being an integer also implies a required integer relationship between fosc and fc.
Another adverse effect of low comparison frequency indirectly affects the switching speed. It is well known in the art that the switching speed is inversely proportional to the loop bandwidth LBW. It is often the case that the LBW must be substantially lower than the comparison frequency, since the signals with the energy at the comparison frequency may occur in the loop, due to the leakage of this frequency from the phase comparator. To prevent phase modulation of the oscillator and resulting discrete sidebands of the oscillator, the loop must attenuate the undesired energy at fc frequency, and to do so it may be necessary to reduce the loop bandwidth far below the fc frequency. While reducing the LBW may eliminate the unwanted sidebands, it will also reduce the switching speed of the synthesizer. Reduced LBW may also reduce the rejection of the internal oscillator phase noise, which is otherwise eliminated by the loop. To address the switching speed, many synthesizer designs often include dynamically switched LBW—a wide one and a narrow one. The wide bandwith is selected during acquisition stage to speed up the acquisition, upon which a narrow bandwidth is switched-in for normal operation. While in this way the switching speed is improved, the oscillator noise rejection remains poor with narrow LBW.
Another adverse effect associated with narrow LBW is related to phase modulation of the oscillator due to mechanical shock or vibration (referred to as microphonics). In general, mechanical vibration can cause parametric modulation of various components of a PLL (such as varactor diodes, inductors, etc.), which can result in phase modulation of the oscillator. The loop will reject the spectral energy of the mechanical vibration within the LBW, but outside the LBW, the loop will not be able to protect the oscillator of being phase modulated by the vibration. Therefore, wider loops are more resistant to shock and vibration, and narrower loops less.
Returning back to the discussion on the available types of PLL synthesizers, it is important to note that the solution for low cost, small size and low power consumption in modern, consumer type communications system relies on the ability to integrate the entire synthesizer, short of the oscillator, on a single integrated circuit, i.e. the application often requires the so called single-chip solution. Some more advanced applications are demanding the integration of the oscillator, too, on the chip.
There are three types of the single loop PLL synthesizers presently used as a single-chip low cost solutions in the art, and they primarily differ from each other in the way the Ntot division ratio is accomplished. The three types are the following: the fixed modulus prescaler type, the dual modulus prescaler type and the fractional—N type PLL synthesizer.
The fixed modulus prescaler contains a fixed prescaler with division ratio P, where Ntot=P·N and N is a programmable counter division ratio. Per equation (3), the minimum increment is limited to P·fc. As an example, if P=10 and step size is to be 30 kHz, the comparison frequency fc needs to be as low as 3 kHz, which will cause an increase of noise gain in the loop by another 20 dB in the previous example. Clearly, the limitation of this type of the synthesizer only aggravates the issues discussed above, and in many applications it can not provide the satisfactory performance.
The dual modulus prescaler type offers a division ratio of Ntot=N·P+A, such that the output frequency is defined by:fosc=(N·P+A)·fc  (4)
The minimum increment with this type of synthesizer is equal to comparison frequency Δfosc=fstep=fc, since A is an integer which can be incremented by 1. This synthesizer operates on the principle of dual modulus prescaling, whereby the fosc is first divided by (P+1) for A cycles of the divided signal, and then the modulus is switched to P, for subsequent (N−A) pulses of the divided waveform. The total pulse count in one full division cycle is therefore A·(P+1)+(N−A)·P=N·P+A. This type of synthesizer is widely used and presents the “work horse” of the industry.
The fractional—N type synthesizer (FNS) represents the current state of the art in the field of a single-chip PLL frequency synthesis. IT is the only type that offers non-integer division ratios, and therefore offers higher comparison frequency than the step size. Currently the available FNS integrated circuits offer sub-integer ratios of 1/5 and 1/8 allowing for output frequency increments equal to fc/5 or fc/8. The increased comparison frequency by a factor of 8 will reduce the noise multiplication figure by 18 dB, but nevertheless, the multiplication factor will still remain relatively high.
The FNS presents an augmentation of the dual modulus prescaler synthesizers. The capability to divide with non-integer numbers is effectively accomplished by varying the total division ratio as function of time, whereby the total division ratio is not held constant at Ntot, but rather it is dynamically changed in time from Ntot to Ntot+1 and vice versa, in a controlled way, such that an average division ratio that is not an integer, but rather a fractional number is obtained. The principle of operation of fractional—N synthesizer can be better understood by referring to FIG. 1, where a functional block diagram of an FNS is illustrated. As shown in this figure, the augmentation portion that is responsible for the extension of the regular dual modulus prescaler synthesizer into a fractional—N type synthesizer contains a phase accumulator with modulo M division ratio, a fraction register F and a compensation DAC. Modulo M phase accumulator counts pulses as they arrive from the N counter and effectively divides this pulse rate by a factor equal to M. Normally, the systems divides by Ntot=N·P+A, in the same way as regular dual modulus type. However, every time the M counter reaches a value equal to F, which is a value preset by a fraction register F, the M counter issues one overflow pulse that sets the dual modulus prescaler to divide by (P+1) for one extra cycle of the divided signal, effectively increasing A to A+1. The system then starts dividing by Ntot+1=N·P+A+1, but only until the next pulse from the N counter, which restores the divide by (P+1) cycle back to A, at which time the system switches back to normal division rate of N·P+A. The system continues to count at this normal rate until the next overflow pulse from the M counter, when it switches to Ntot+1 rate, and the whole cycle repeats again.
The output frequency of fractional—N synthesizer is given by:                     fosc        =                              (                                          N                ·                P                            +              A              +                              F                M                                      )                    ·          fc                                    (        5        )            where M is the modulus of the phase accumulator and F is the fraction register value. The frequency increments of Δfosc=fstep=fc/M are available. Varying F from 0 to M−1, contiguous steps are possible.
Examining the division rate as a function of time, it becomes clear that only the average frequency at the output of the programmable N counter equals that of the comparison frequency, while the instantaneous frequency (i.e. the instantaneous pulse rate at the N counter output) is never equal to the comparison frequency fc, which has a constant, uniform pulse rate. The N counter output pulse rate alternates in time between the two values, one equal to fosc/(Ntot1) and the other to fosc/Ntot. The phase/frequency detector PFD sees on one side a uniform pulse rate fc coming from the reference divider R, and on the other side a variable frequency coming from the N counter. Since these two rates are never the same, the phase detector at the end of each comparison cycle always issues an error signal, equal to the phase discrepancy of the two signals at the end of that comparison cycle. This error signal needs to be completely removed from the loop, otherwise severe phase modulation of the oscillator would occur (it can be shown that the accumulated phase error in one complete fractional cycle amounts to 360° at the oscillator frequency). The removal of the phase error signal is accomplished by the means of the compensation DAC which injects correction signals directly into the loop. On a cycle by cycle basis, the phase error is substantially predictable and it can be precisely determined as a function of count state of the M counter. The compensation DAC can therefore be directly driven by the M counter. At each comparison cycle the DAC generates a precise charge matched to the instantaneous phase error and attempts to cancel this phase error. The required precision of the correction charge is substantial. As an example, for 60 dB sideband purity of the oscillator, matching of the correction pulses and phase error in the order of 0.1% is required. Although it is achievable in the state of the art FNS ICs, it is often difficult to maintain this kind of precision over wide frequency range and temperature range since both the compensation DAC and PFD will exhibit some variations due to these factors. This is primarily due to the effect of temperature on propagation delays within the IC in the PFD, as well as effects of both the frequency and temperature on the current sources used in the compensation DACs. For wide tuning range oscillators, it is often necessary to provide an external adjustment for the compensation current of the DACs in the application circuit (outside of the IC), which may complicate the manufacturing process. Due to these factors, the ultimate phase error compensation is limited, and to ensure sufficient spectral purity of the oscillator, the loop filter often must be called upon to assist in further reduction of the sidebands, resulting in the LBW having a cut-off much below the fc/M, i.e. much below the step increment frequency. The reduced LBW, as mention earlier may adversely affect both the switching speed and the rejection of the internal oscillator phase noise. There are efforts in the art of fractional—N synthesis to improve phase error correction methods and alleviate some of the problems discussed above. More advanced approaches include utilization of higher order sigma-delta modulators in the role of a compensation DAC, which should push the phase error energy out of the loop bandwidth, towards higher frequencies by proper noise shaping of the correction pulses.
Finally, it should be mentioned that all of the above discussed prior art synthesizers use a phase-frequency detector which is inherently limited in speed. The type of the PFD used in the prior art relies on flip-flops with reset line fed back from the output. The propagation delay and set-up and hold times of this circuit limits the maximum speed (or frequency) of the PFD operation. Further more, in phase lock condition, there is a well known phenomena called a “dead zone”. This is a point where phase detector gain goes to zero, which can cause instability in the loop, resulting in random phase fluctuation (low frequency phase noise) of the oscillator.
Thus, those of skill in the art will recognize the need for an alternative solution in the PLL frequency synthesis, one that can operate at much higher comparison frequencies, thus substantially improving phase noise performance, and at the same time providing fine frequency step increments of the output oscillator frequency.